domingo, 30 de mayo de 2010

FPGA signal processing for radar/sonar applications

Radar and sonar applications are signal-processing intensive and heavily rely on the efficient implementation of such digital signal-processing (DSP) algorithms as filtering, transforms and modulation. In past systems, conventional digital signal processors were used to perform many of these algorithms. However, field-programmable gate arrays (FPGAs) deliver an order of magnitude higher performance than traditional DSPs. A key reason is that an FPGA can side step the classic Von Neumann architecture's instruction — fetch, load/store bottleneck — found in most DSPs. Another reason is the FPGA's lower power consumption.

When approaching the problem of implementing signal-processing functions within an FPGA, designers have developed the mindset that these functions must be optimally coded from the ground up for their application or significantly modified. However, silicon-optimized, high-precision math functions are being developed for specific applications as part of the programmable logic product offering of many vendors, including Altera, making complex systems easier to manage and lower risk.


Changing requirements in radar applications
Modern military radar systems have evolving requirements, both in how the systems are designed and how the end user uses the data. This results in some of the same design changes in electronic systems affecting both the military and commercial design communities. That is, the need for smaller, energy-efficient systems with high processing-power requirements. This makes low power consumption a key driver in most designs.

With warfare having become more urban, ground clutter and background noise take on additional significance for the radar operator, thus demanding more processing power and better algorithms. Overlaying data from multiple sensors and known terrain features is one approach to increasing resolution, but this too has impacts on system-processing requirements and user-decision models.

High-speed digital systems make new digital beam-forming technologies possible, increasing the number of beams and nulls available for warfighting and surveillance missions. More digital logic also allows designers to make early decisions on actionable intelligence and to meta-tag sensor data earlier for more efficient analysis. These and other emerging techniques will allow for the creation of better radar or sonar systems, but each requires additional signal-processing resources.

One of these resources is the emerging class of high-performance FPGAs. One of the primary differences in the past between FPGAs and application-specific integrated circuits (ASICs) has been greater complexity in the latter class of devices. However, with the 65 nm generation of FPGAs and 45 nm devices on the horizon, FPGAs in sensor systems have become nearly as complex as ASICs. This complexity comes from rapidly increasing logic density, as well as from the integration of the many different processing functions now integrated into one device.

Modern approaches to radar DSP
As FPGAs increase in density and performance capability, more signal-processing functions can be incorporated and migrated to the front end containing the exciter/receiver of the radar (or sonar) system. This may include waveform generation, filtering, matrix-inverse operations, and signal correlation.

A representative multi-element radar element is shown in Figure 1, with multiple signal-processing and beam-forming elements represented in a single logic element. The design of this single FPGA quickly becomes complex, particularly if the beam-forming algorithms allow for multiple beams and nulls in the active array.

An all-FPGA design for signal processing
Fitting multiple DSP functions into a single FPGA has many integration challenges, but also offers significant advantages to the designer in performance and flexibility.

The primary reasons for integrating DSP functions into a single FPGA are system-level reductions in size, weight and power. For example, eliminating the transfer pathways between separate FPGAs and DSPs significantly reduces power consumption and, therefore, heat. This, in turn, reduces the system-cooling burden of the design. Recent releases of design and place-and-route software, such as Altera's Quartus II design suite, have advanced power-awareness features that significantly reduce dynamic power use of the FPGA. These options can be important to the designer; the benchmark of device logic density among competitive FPGA providers is beginning to give way to functionality-per-watt metrics, due to the sensitivity of power and cooling requirements in emerging systems.

Performance is also a key driver as FPGA-pipelined signal processing has become more reliable and faster than traditional processing technologies. In applications where performance is the driving parameter, efficiency can be sacrificed for application speed, where a memory-intensive, massively parallel floating-point math operation is desired. Alternatively, highly iterative DSP calculations can be implemented for applications where moderate performance is allowable, but where logic-element usage is limited.

This leads to the advantage of flexibility. The designer has the flexibility to decide between high-speed performance and the number of logic elements in every DSP operation, whereas calculation bandwidths and iterations would be more difficult and costly to modify in a dedicated DSP device. In addition, consolidating DSP functions within an FPGA allows for post-design system changes in the signal-processing architecture, whereas using separate DSPs locks the designer into a fixed set of chip interfaces once the board is designed. FPGA designers can alternately switch between 9-bit, 18-bit or 36-bit or 18-bit complex math functions without changing the system hardware.

Additional flexibility can be designed into the system when the designer uses fast-embedded processors for the execution or routing of complex floating-point operations. These functions are useful for radar applications.

FPGA DSP functions in radar/sonar applications
Several DSP functions are needed for radar or sonar processing near the receiver element. Each function should be closely examined to determine whether the application will show substantial speed and performance improvements through implementation in an FPGA. In some cases, these operations can be efficiently implemented using an FPGA embedded processor, even for highly complex and adaptive operations.

When a radar or sonar application calls for these operations to be performed with floating-point arithmetic, FPGAs have significant flexibility advantages if the design team takes advantage of a strong architecture-based design approach. Large floating-point math operations can be performed in standard logic cells (the least efficient option), in dedicated reduced-instruction-set-computer (RISC) embedded processors (the most flexible option), or in dedicated floating-point multiplier logic (the most efficient option).

FPGA providers and third-party developers offer efficient and accurate floating-point operators, Fourier-transform tools and filter compilers to FPGA designers as intellectual property (IP). Engineers should conduct their own research on the current availability of advanced DSP functions, but a great deal of preliminary information can be obtained through the technical representatives of programmable logic device (PLD) vendors.

Digital up/digital down signal conversion
The upconversion and downconversion of high-frequency signals are experiencing a dual migration, into the digital domain, and into the same monolithic device (either the ASIC or FPGA) that performs the baseband processing. This push toward more digital, software-radio-style signal-processing techniques provides significant advantages to the system in signal accuracy and speed.

The closer to the RF front end (or the acoustic transceiver front end in sonar systems) that signals can be digitized, the fewer the analog-signal vulnerabilities that are introduced to the system. This includes high-order mixing products, error-vector-magnitude (EVM) impairments due to phase/magnitude imbalance, carrier feed-through, harmonics, and sideband noise.

More important than signal integrity, however, is the design flexibility that the digital domain allows the radar-system designer. Dynamic filtering and conditional signal-processing algorithms significantly improve performance, as well as reduce implementation losses and the time required for the design cycle. While these advantages involve trade offs between power consumption and digital bandwidth, modern FPGAs provide designers much greater flexibility in mitigating power consumption, including the support of selectable core voltages, or critical-path power analyses.

The greater the numbers of on-chip resources available in FPGAs, the more designers are enabled to incorporate polyphase filtering and downconversion in the digital domain, as reflected in Figure 3. Multiple onboard or external numerically controlled oscillators (NCOs) can allow very high phase discrimination with high-capacity FPGA devices. This application is useful for prototyping, research and development, where designers can incorporate and test multiple-phase resolutions without significant hardware investments by using hardware-in-the-loop test methodologies.

Algorithmic functions
Examples of algorithmic math functions in radar systems include recursive least-square and square-root operations. Many designers have implemented these functions in C-based processors (in fixed-decimal and floating-point operations), or with proprietary FPGA VHDL operations. The current generation of FPGA devices include embedded processor and logic-cell resources to efficiently implement these processes; future generations will also have these capabilities. Additionally, IP cores and reference designs are becoming available to transition anywhere from dozens to hundreds of these operations into a single FPGA.

Tools are available to translate processor-based algorithms from C code to hardware languages, such as very high-level descriptive language (VHDL). These tools can be used to optimize certain logic functions from a standard main processor into an FPGA co-processor operating in parallel with the main processor, or to move entire operations from the main processor to the FPGA hardware. This provides an additional dimension of flexibility to the radar- or sonar-architecture designer's toolkit.

Complex matrix inversion
Matrix inversion is an important element of adaptive-array designs and standard spatial-transceiver-array processing (STAP). These operations are commonly performed in fixed hardware elements, though efficiently implemented embedded processing has been demonstrated in some radar/sonar development programs.

The logic-element size and potential parallelism of a matrix inversion engine depends on the size of the array used in the radar system. As the size of the array is increased, so does the number of floating-point multiplications required by the system. Therefore, in larger arrays, there are more trade-off options between the speed of the system and the number of logic elements required by the system (both of which increase as the parallelization of the architecture increases).

Implementing this function using a combination of a DSP and a group of internal memory blocks is the most likely design path for radar-system designers. As these operations are often tailored to the adaptive-array algorithms of the radar system, they are likely to be custom designed in VHDL. However, reference designs that are optimized for the place-and-route capabilities of an FPGA device can be offered or designed to order from the FPGA manufacturer, if required for the radar or sonar system.

Fast-Fourier transforms
The bandwidths of many systems, including radar/sonar and test/measurement systems, are beginning to exceed the capabilities of dedicated DSPs. Implementing fast-Fourier transforms (FFTs) and their inverses in FPGA logic has advantages in prototyping and scalability, and offers design flexibility between a system's speed and the number of required logic elements. For example, massively parallel implementations can be designed and distributed among the logic elements of a single or multiple FPGAs. However, while these implementations can significantly reduce latency, they impose the penalty of a greater number of logic elements.

In fact, the primary flexibility advantage of an FPGA for FFTs is the ability to select the optimal balance between these two parameters in the initial design. This is fortunate, because the implementation of large or complex FFTs should be the primary factor in any design, and the advantages of an FFT implementation in an FPGA (Figure 4) are apparent. However, creating code or modifying existing code from previous designs can be cumbersome when testing and verifying code units. Therefore, what is needed is a comprehensive suite of FFT design tools that allows a nearly infinitely scalable FFT design. These tools should allow scripted logic distribution among multiple FPGAs where necessary. They should also be able to automatically generate numerical coefficients having floating-point accuracy. Customer inputs are being taken now for such tools.

Because radar, sonar and digital-communication system designers must focus on the complications of multi-element beam-forming and waveform generation — not FFT design — programmable logic vendors such as Altera have internal tools and generators for conducting large, difficult element transformations. This includes reference designs and core IP wizards for standard and non-standard designs, as well as FFT co-processors, which are important design aids in the programmable logic offering.

Design flow
DSP logic designs are commonly executed from an initial model in simulation languages, such as Matlab or Simulink. These models are the most common, but not the only sources for designers to access optimized DSP IP offered through FPGA providers. The linkage between modeling and hardware implementation is important, not only for design simplicity, but for simulation and verification against the model.

As the design density for FPGA-based sensor systems increase, full system modeling and simulation will become more time consuming. Compile, simulation, and place- and-route times will increasingly become discriminators when selecting FPGA and design-software vendors. Furthermore, multiprocessor and distributed processing options for design software will be necessary to keep up with design complexity.

To cope with these trends, and to achieve the greatest signal-processing performance in their sonar or radar systems, designers are encouraged to consider options beyond their own VHDL modules or other internally developed IP. Specifically, they should consider working with programmable logic manufacturers to develop tailored DSP cores, or find ways to improve and optimize their designs through advanced place-and-route methods available for FPGA design tools. This is because the advanced capabilities of integrated circuits enabled by increasingly sophisticated fabrication technologies cannot be fully harnessed without flexible and effective design techniques.

Jorge L Polentino U
19769972
CRF
http://mobiledevdesign.com/hardware_news/fpga-signal-processing-sonar-dsp-1207/

MEMS: Diseño de un microrelay realizado con tecnología SOI

El desarrollo de dispositivos MEMS (Micro Electro-Mechanical Systems) ha experimentado un continuo crecimiento con nuevas áreas de aplicación. Paralelamente, la tecnología SOI (Silicon On Insulator) a demostrado ser una interesante opción para ser utilizada en la fabricación de microsensores y MEMS surgiendo en el mercado más opciones de servicios de fabricación en este tipo de tecnología. En este trabajo se presenta el diseño de un Demostrador con tecnología MEMS sobre obleas de tipo SOI para ser fabricado por la empresa Tronic's a través del sistema Europractice.

La industria de dispositivos MEMS sigue experimentando un gran crecimiento y ampliando cada
vez más sus campos de aplicación. Así mismo, lasexigencias del mercado hacen que crezca la demanda de dispositivos de mayor rendimiento y confiabilidad. La utilización de SOI como substrato para la fabricación de dispositivos MEMS a demostrado ser extremadamente versátil [1]. Este material tiene grandes ventajas respecto del silicio policristalino [2]. Sus principales características son el reducido “stress” residual y la posibilidad de combinar circuitos eléctricos con elementos micromecánicos. También se caracteriza por su gran resistencia en ambientes extremos y corrosivos, soportando altas temperaturas y niveles de radiación. Desde el punto de vista del diseño y la producción de MEMS, tiene además la ventaja de requerir un menor número de máscaras para el proceso de fabricación. Como consecuencia de lo expresado surgen en el mercado más alternativas para la fabricación de MEMS utilizando obleas de tipo SOI. Si a eso se le suma la posibilidad de obtener prototipos a bajo costo mediante el sistema de servicios MPW (Multi Project Wafer) la utilización de SOI para el desarrollo de MEMS se convierte en una opción viable e interesante.
En este trabajo se presenta el diseño de un Demostrador con tecnología MEMS sobre obleas de tipo SOI para ser fabricado mediante el uso del servicio MPW [3], para la fabricación de prototipos, que la empresa Tronic's ofrece a través del sistema Europractice.

DISEÑO DEL DEMOSTRADOR
Se diseñó un Demostrador de MEMS conteniendo veintidós variantes de una unidad microrelay tomada como referencia. A partir de la estructura base del microrelay se diseñaron varios dispositivos con diferentes dimensiones y/o estructuras con el objeto de analizar su comportamiento mecánico y eléctrico. El dispositivo de referencia es un microrelay de contactos laterales accionado mediante actuadores electrostáticos de accionamiento lateral. Los actuadores están formados por dos peines, uno fijo y el otro móvil sujeto a un anclaje por medio de un resorte. El accionamiento electrostático presenta ciertas ventajas respecto a otros métodos alternativos. Por su parte la actuación lateral resuelve algunos de los inconvenientes que presenta el accionamiento vertical, como por ejemplo la alinealidad de la fuerza de actuación respecto al desplazamiento [4] [5]. El actuador de tipo peine provee una fuerza electrostática lineal [6] [7], producida por la aplicación de una diferencia de potencial, que depen de en granmedida de las dimensiones del actuador [5]. Un parámetro importante es la tensión que se debe aplicar para producir el cierre de los contactos del microrelay.
Se determinó que el aumento en el ancho de los dedos del peine, si bien logra una disminución de dicha tensión, su influencia no es muy significativa comparada con la de otros parámetros, por lo cual se mantuvo constante salvo en uno de los dispositivos solo a los fines de verificar los cálculos realizados. Los parámetros que se tomaron en consideración para el diseño de los diferentes dispositivos del Demostrador son la separación entre dedos del peine, la separación en el extremo de los dedos del peine, la cantidad de dedos y la longitud y espesor del elemento elástico. También se utilizó una estructura alternativa para éste último para evaluar su comportamiento. El tamaño del Demostrador es de 3400x3100 μm y los microrelay dentro de dicha área van desde 500x400 μm a 900x400 μm.




PROCESO SOI
El proceso de fabricación SOI utilizado [3], si bien por un lado presentó limitaciones en cuanto a la flexibilidad en el diseño, por otro lado simplificó las tareas, ya que solo fue necesario el diseño de una máscara, correspondiente a la estructura de silicio del conjunto. Las restricciones impuestas por el proceso SOI impidieron obtener la metalización de los contactos laterales del microrelay, por lo que se realizará posteriormente mediante la técnica de evaporación. Para ello se removerá el encapsulado del Demostrador colocado en la etapa final del proceso de fabricación. Se dispusieron, alrededor del área del Demostrador, una serie de pads a los cuales se conectan seis microrelay con el objetivo de ensayar algún tipo de encapsulado posterior. El tamaño de los pads es de 200x200 μm en uno de los laterales y 200x320 μm en el otro con un pitch de 240 μm en ambos casos. La capa estructural de silicio, que forma la estructura móvil de los dispositivos, es de 20 μm de espesor y la capa de óxido de la oblea SOI de 0.4 μm.

DISEÑO DEL LAYOUT
El diseño del Demostrador fue realizado en el IMEC de Leuven-Bélgica ajustándose a las reglas del proceso de fabricación. Para el diseño de la máscara se utilizó el editor de layout de Cadence, el cual está estructurado en forma jerárquica en varios niveles. El nivel más bajo lo constituyen una serie de celdas con estructuras básicas definidas en forma paramétrica para facilitar la construcción de los diferentes dispositivos del Demostrador que difieren básicamente en sus dimensiones. Las celdas definidas corresponden a las estructuras de los peines de los actuadores, a los contactos del microrelay, a los pads de conexionado, a los anclajes de la estructura y a las vigas del resorte. La posibilidad del editor de trabajar con este tipo de celdas facilitó mucho el diseño del layout. El Demostrador diseñado se envió para su fabricación y se prevé realizar en IMEC la caracterización de los prototipos para evaluar sus resultados.
Se diseñó un Demostrador de dispositivos MEMS con tecnología SOI utilizando uno de los procesos de fabricación disponibles en el mercado. Se incluyeron en el Demostrador varios microrelay con diferentes estructuras y dimensiones para ensayar sus características eléctricas y mecánicas. Se diseñó una biblioteca de celdas paramétricas para el diseño del layout. El diseño completo del layout requirió solamente una máscara para su implementación.
Jorge L Polentino U
19769972
CRF
http://www.iberchip.org/VIII/docs/posters/p27.pdf

MEMS-Switched Reconfigurable Antennas

The integration and use of RF MEMS switches in microstrip patch antennas and feed structures were investigated for developing reconfigurable multi-band antennas. The current application focuses on the development of a dual L/X-band antenna that would support several satellite or UAV-based communications and radar applications such as SAR, terrain mapping, GMTI, AMTI, etc. A reconfigurable patch module (RPM) was designed and fabricated consisting of a 3x3 array of patches connected together using MEMS switches (we simulated MEMS switches with ideal OPEN/CLOSED switches). This RPM could be used as a basic element in a tile/conformal architecture. Stripline power dividers and blind via transitions were developed to demonstrate feed structures that could be located below the radiating aperture.

Reconfigurable multi-band phased-array antennas are receiving a lot of attention lately due to the emergence of RF MEMS (micro-electro-mechanical systems) switches [1-6]. A MEMS-switched reconfigurable multi-band antenna is one that can be dynamically reconfigured within a few microseconds to serve different applications at drastically different frequency bands, such as communications at L-band (1-2 GHz) and synthetic aperture radar (SAR) at X-band (8-12.5 GHz).
The Air Force also uses both ground- and airborne- moving target indication (GMTI/AMTI) at these frequencies in order to detect moving targets such as vehicles on the ground and low observables in the air. The RF MEMS switch is attractive because it shows of achieving excellent
switching characteristics [2] over an extremely wide band (DC-40 GHz and upwards). These switches can also be used to develop wideband phase shifters [3]. Although there is currently a tremendous amount of research in RF MEMS devices, reliability and packaging of the switches continue to be problematic. The switches are also limited in their power handling capability.In this work, we do not focus on the development of the MEMS switches themselves. Rather, we want to use them as control elements in a reconfigurable antenna. Since the actual MEMS switches were not available to us at the time of this work, we simulated the MEMS switches using ideal OPEN and CLOSED circuits. There is actually a great deal of work to develop optimal radiating elements and feed structures in order to achieve the desired multi-band performance.
Reconfigurable Patch Module (RPM): We investigated the design and fabrication of a dual L/X-band reconfigurable antenna. Microstrip antenna elements were chosen due to their inherent lowprofile, which is suitable for satellite and UAV applications. We used RT/duroid 5880 material with a dielectric constant of 2.2 and a loss tangent of 0.0009 at 10 GHz. Two material thicknesses were investigated: 0.062” and 0.125”. The material thickness must be chosen carefully, since it controls both the bandwidth and array scanning performance. The thicker the material, the more bandwidth, particularly at the low frequency end. However, if the substrate becomes too thick, surface waves are generated and array scanning performance and efficiency is lost. Figure 1 shows a picture of the 3x3 RPM fabricated on 0.125” duroid. The patches are 0.370” square and separated by 0.590” on center. Interconnecting tabs are 0.050” wide and 0.085” long. The “reconfigurable” antenna was actually fabricated as two separated prototypes (OPEN and CLOSED configurations) for testing in the laboratory.






shows the measured return loss for the 3x3 RPM in Figure 1 for both the CLOSED (L-band) and OPEN (S-band) configurations. We were able to achieve 1.2% impedance bandwidth for the L band configuration and greater than 7% bandwidth at X-band. This bandwidth was limited rimarily by the substrate thickness. Figures 3 and 4 show the measured radiation patterns at both L-band and X-band. Computer simulation results for both the return loss and radiation patterns agree well with the measurements, and will be shown in the talk.



Jorge L Polentino U

19769972

CRF
http://www.appliedradar.com/Papers/aps01_mems.pdf

SISTEMAS MICRO-ELECTRO-MECÁNICOS PARA DETECCIÓN DE GASES. Sensores avanzados para múltiples aplicaciones

Los Sistemas Micro-Electro-Mecánicos (Micro-electro-mechanical systems (MEMS) )y de tecnologías de película delgada permiten la integración de circuitos electrónicos y matrices de sensores multifuncionales fabricados en sustrato de silicio como detectores químicos, mecánicos y de parámetros físicos. También la integración ofrece la posibilidad de acortar el camino entre los sensores y las técnicas de reconocimiento. Es importante fijarse que los sensores biológicos tienen igualmente la facultad de funcionar como sensores múltiples por ejemplo el sentido del tacto agrupa sensaciones de temperatura, presión, viscosidad de líquidos, además de disponer de un mecanismo de retroalimentación y de memoria.

Los MEMS son sensores avanzados para detectar simultáneamente varios parámetros, temperatura, presión, radiación, gas y concentración de vapor, olor, aceleración, inercia, campos eléctricos y magnéticos y muchos más, proporcionan no solamente alta relación señal ruido en un gran rango dinámico sino que también presentan buena sensibilidad. En general, los sensores consisten en dos elementos: un detector y una plataforma que comunica con el detector a través de un interface activo con variables eléctricas, mecánicas, ópticas o impedancia química. La plataforma debe permitir generar la salida de señales eléctricas que transportan la información proporcionada por el detector.

El principal elemento utilizado en los MEMS es el silicio. La tecnología usada en la fabricación electrónica de las micromáquinas es cualquiera de las utilizadas en la fabricación de circuitos integrados, la fotoeléctrica, difusión ,oxidación , etc. La técnica como silicon-to-silicon y silicon-to-glass es la que se suele emplear permitiendo la realización de sensores de olor.

Sensor de superficie acustica SAW
los sensores mas comunes estan formados por dispositivos piezoelectronicos. la figura muestra la estructura basica de un sensor acustico electronico de gas. este tipo de sensores se utiliza para la realizacion de sensores multifuncionales fisico y quimicos; esta estructura se utiliza para sensores microscopicos de viscosidad, humedad, detectores de humo, sensores de gas, y sensores de campo magnetico-electrico normalmente el deposito de sustancias en la pelicula delgada se utilizan parala medida de estos parametros al producir cambios fisicos y quimicos que hacen variar la frecuencia de resonancia de la pelicula. los cambios en los sensores basados en el desplazamiento de frecuencia de resonancia, son producidos por causas mecanicas, quimicas u otras perturbaciones



Si el resonador se pone junto con una película delgada, las condiciones de funcionamiento varían. Una película dieléctrica modifica su funcionamiento bajo condiciones mecánicas, mientras una película conductora modifica ambos su funcionamiento eléctrico y mecánico. Las perturbaciones mecánicas y eléctricas causan desplazamientos de la frecuencia de resonancia. Si asumimos que la cantidad de corriente interna es aproximadamente y , donde F es el potencial eléctrico, y ho es la densidad de carga eléctrica, los cambios en el funcionamiento y desplazamiento de frecuencia de resonancia vienen dados por:



donde U es la energía acústica cargada en modo resonador, T es la tensión de esfuerzo, y * indica
un conjugado complejo. Las condiciones de la interface son:



Donde H es el campo magnético relativo a al campo eléctrico a través de la ecuación de Maxwell. El régimen de la oscilación del resonador piezoeléctrico puede ser modificado mecánica o eléctricamente. Las perturbaciones eléctricas pueden ocurrir en la película metálica con diferentes valores de conductividad en el resonador o si el resonador se introduce en un electrolito de conducción iónica. La influencia mecánica, química y eléctrica en sólidos y fluidos en la superficie del sensor depende de la interface entre el resonador de cuarzo y la resonancia. Algunos efectos en líquidos y sólidos hacen oscilar el resonador y modificar la resolución del sensor. La resolución del sensor se determina por la respuesta en el desplazamiento de la frecuencia de resonancia, a perturbaciones y la capacidad de monitorizar los cambios en desplazamiento de frecuencia. Cuando un resonador de cuarzo libre se pone en contacto con un sólido o fluido, parte de la energía acústica se trasmite fuera del resonador.

El acoplamiento acústico define el desplazamiento de la frecuencia de resonancia, y modifica el factor Q de calidad.

La siguiente figura muestra la superficie de una plataforma de onda acústica usando varios sensores de película delgada depositados en un resonador piezoeléctrico. En una mezcla de gases, cada película detecta un componente determinado.



La figura muestra otra superficie de onda acústica (SAW) configurada con silicio, en un sustrato no piezoeléctrico. El traductor interdigital esta construido por ZnO, un material piezoeléctrico. La línea SAW es parte de un circuito oscilador. Cuando la sensibilidad cambia los parámetros mecánicos también lo que se produce por la presencia de un gas cuya presión desplaza la frecuencia de resonancia y cambia la velocidad de propagación en el SAW. La línea de referencia del SAW, construida con película de cristal pasivo se usa para calibraciones.



Sensores Electroquímicos
Otra familia de sensores multifunción son los sensores de gas electroquímicos ver Figura , que se usan con celdas galvánicas en estado sólido para media de presiones parciales de gases como CO2, NOx, SOx, y gases de hidrocarburos. El sensor opera como una batería: la fuerza electromotriz cambia la función química modificando el cátodo en presencia de gas de acuerdo a la siguiente reacción:




Pt electrode, gas gas sensitive film ion conductor reference electrode Pt electrode


Por ejemplo, para medir la f.e.m. y la presión parcial del NO2 una celda galvánica, produce:


La reacción en la celda es:

La relación ente la f.e.m. y la presión parcial de NO2 viene dada por:

Donde G es la energía de Gibbs, P es la presión parcial, k es la constante de Boltzmann, y T la temperatura absoluta.




Las siguiente figura muestra un sensor de gas para medida parcial de presencia de CO2 fabricado usando tecnología de película delgada. El sensor opera a 350°C y toda su estructura se construye encima de una oblea de silicio con platina de película delgada.















utilizando varios sensores sobre un mismo sustrato de silicio con tecnología MEMS, sólo se necesita un electrodo diferente para cada una de las salidas del sensor.













Jorge L Polentino U
19769972
CRF


MEMS, las nanomáquinas que cambiarán al mundo

La electrónica de consumo ha llegado al estado en que se encuentra hoy gracias a la miniaturización. Sin ella, sería imposible crear circuitos integrados con millones de transistores y un tamaño de solo una fracción de centímetro cuadrado. Sin la microelectrónica, el equivalente de un microprocesador como el que tiene tu ordenador ocuparía el volumen de un edificio de 12 o 14 pisos. No habría iPods ni teléfonos móviles.
Sin embargo, y a pesar de los logros obtenidos en la reducción de tamaño de los componentes electrónicos, los sistemas mecánicos aun requieren de piezas cuyo tamaño es varios órdenes de magnitud más grandes que sus contrapartes electrónicas. Cualquier pieza de un reloj mecánico, por ejemplo, es millones de veces más grande que uno de los transistores integrados en un microprocesador. Pero esta situación está cambiando.
La miniaturización de máquinas electromecánicas ha dado lugar a los MEMS, que silenciosamente han ocupado un lugar en nuestra vida cotidiana. De hecho, el dispositivo capaz de medir la aceleración a la que sometes el mando de tu Wii (un acelerómetro) es un MEMS. Se trata del mismo dispositivo que, instalado en el airbag de un coche determina el momento justo en que se produce un choque y dispara el mecanismo de inflado de las bolsas.

Pero si bien los acelerómetros son quizás los dispositivos basados en MEMS mas difundidos, no son los únicos. Existen sensores de presión, de temperatura y de humedad construidos a partir de piezas que tienen un tamaño similar al de un glóbulo rojo. Forman parte del sistema de control de los más modernos marcapasos, censando la actividad física del paciente para modificar su ritmo cardíaco. También se emplean MEMS en los cabezales de las impresoras de inyección de tinta, como parte del dispositivo que produce la evaporación controlada de la tinta en el momento justo.

Por lo general, estos mecanismos tienen un tamaño mayor al micrómetro (millonésima de metro) y menor al milímetro. Lo que los hace tan particulares es que, a estas escalas, el comportamiento físico que rige a las maquinas convencionales no siempre funciona como la intuición puede indicar. Efectivamente, el incremento en la relación entre la superficie y el volumen de las piezas de un MEMS hace que los efectos electrostáticos y térmicos predominen sobre la inercia o la masa térmica.
Para fabricar las pequeñas piezas que conforman estas maquinas se utiliza una tecnología que, en esencia, es la misma que la empleada para la fabricación de los circuitos integrados. La posibilidad de “integrar” piezas móviles es lo que ha hecho posibles maquinas a escala nanométricas. Existen motores a vapor del tamaño de un grano de polen, engranajes y palancas cuyo tamaño de mide en diámetros atómicos, y hasta pequeños espejos montados sobre soportes móviles, con un tamaño mucho menor al diámetro de un cabello, capaces de enfocar o corregir una imagen.
Los MEMS permiten cada día la creación de dispositivos sorprendentes. Por ejemplo, para evitar la falsificación de una firma, es posible incorporar acelerómetros en una lapicera, para que además de escribir sea capaz de registrar las velocidades y aceleraciones que le imprimió la mano mientras se firmaba. Esto hace prácticamente imposible una falsificación.

Dentro de poco, será factible la fabricación de un dispositivo, que ubicado en el cuerpo de un paciente, analice su sangre y que, en función de los resultados, inyecte los fármacos necesarios en las dosis adecuadas. En caso de ser necesario, hasta podría enviar una señal de alerta para que el paciente fuera atendido de urgencia. Estas máquinas funcionarán como pequeños robots, capaces de realizar tareas que resultan imposibles a una escala mayor.

Se trata de una ciencia que, a pesar de habernos brindado ya una cantidad de soluciones concretas a problemas de ingeniería, recién está naciendo. Pero tiene el potencial de, como decíamos al comienzo, cambiar el mundo.











acelerometro comercial
Jorge L Polentino U.
19769972
CRF

RF MEMS SWITCHES: STATUS OF THE TECHNOLOGY

This paper presents the latest accomplishments in RF MEMS switches, and at the same time, an assessment of their potential applications in defense and commercial systems. It is seen that RF MEMS devices offer spectacular performance at microwave frequencies,but suffer from reliability problems and the potential of relatively high-cost hermetic packaging. Still, this technology offers such tremendous advantages over GaAs and silicon switching devices that, in the author’s opinion, it will find many applications in satellite, base-station and defense applications, particularly at high microwave frequencies.

PROS AND CONS OF RF MEMS
SWITCHES

MEMS switches are surface-micromachined devices which use a mechanical movement to achieve a short circuit or an open circuit in the RF transmission-line (Figs. 1-2). RF MEMS switches are the specific micromechanical switches which are designed to operate at RF to mm-wave frequencies (0.1 to 100 GHz). The advantages of MEMS switches over PIN diode or FET switches are [1]: Near-Zero Power Consumption: Electrostatic actuation requires 30-80 V, but does not consume any current, leading to a very low power dissipation (10-100 nJ per switching cycles). On the other hand, thermal magnetic switches consume a lot of current unless they are made to latch in the down-state position once actuated.
Very High Isolation: RF MEMS metal-contact switches are fabricated with air gaps, and therefore, have very low off-state capacitances (2-4 fF) resultingin excellent isolation at 0.1-60GHz. Also, capacitive switches with a capacitance ratio of 60-160 provide excellent isolation from 8-100Hz. Very Low Insertion Loss: RF MEMS metal-contact and capacitive switches have an insertion loss of 0.1dB up to 100GHz. Linearity and Intermodulation Products: MEMS switches are extremely linear devices and therefore re-


sult in very low intermodulation products in switching and tuning operations. Their performance is 30-50 dB better than PIN or FET switches. Potential for Low Cost: RF MEMS switches are fabricated using surface micromachining techniques and can be built on quartz, Pyrex, LTCC, mechanicalgrade high-resistivity silicon or GaAs substrates. RF MEMS switches also have their share of problems, and these are: Relatively Low Speeds: The switching speed of most electrostatic MEMS switches is 2-40 μs, and



High Voltage or High Current Drive: Electrostatic MEMS switches require 30-80 V for reliable operation, and this requires a voltage up-converter chip when used in portable telecommunication systems. Thermal magnetic switches can be actuated using 2-5 V, but require 10-100 mA of actuation current. Power Handling: Most MEMS switches cannot handle more than 200 mW although some switches have shown up to 500 mW power handling (Terravicta and Raytheon). MEMS switches that handle 1-10 W with high reliability simply do not exist today. Reliability: The reliability of mature MEMS switches is 0.1-40 Billion cycles. However, many systems require switches with 20-200 Billion cycles. Also, the long term reliability (years) has not yet been addressed. It is now well known that the capacitive switches are limited by the dielectric charging which occurs in the actuation electrode, while the metalcontact switches are limited by the interface problems between the contact metals, which could be severe under low contact forces (in electrostatic designs, the contact forces are around 40-100 μN per contact).

It is important to note that the reliability and packaging issues have been the limiting factors to the quick deployment of RF MEMS switches, and they are currently under intense investigations. DARPA has initiated two programs in 2002 and 2003 to address these problems, the RF MEMS Improvement program (Dr. Larry Corey), and the HERMIT program (Dr. Clark Nguyen), and it is expected that some of these problems will be solved in the coming 2-3 years. Packaging: MEMS switches need to be packaged in inert atmospheres (Nitrogen, Argon, etc..) and in very low humidity, resulting in hermetic or nearhermetic seals. Hermetic packaging costs are currently relatively high, and the packaging technique itself may adversely affect the reliability of the MEMS switch. Microassembly (Fig. 3) and Analog Devices have both developed excellent packages for RF MEMS switches. The Microassembly package is based on gold-to-gold thermo-compression at 250◦C while the Analog Devices package is based on glass-to-glass seal at 400−450◦C. Other companies which have packaged switches are Terravicta (ceramic package) and Omron (glass-to-glass). Cost: While MEMS switches have the potential of very low cost manufacturing, one must add the cost of the packaging and the high-voltage drive chip. It is therefore hard to beat a $0.3-0.6 single-pole doublethrow 3 V PIN or FET switch, tested, packaged and delivered. It is for this reason that Prof. Rebeiz believes that RF MEMS switches will be first used in defense and high-value commercial applications and not in cellular phones.



DETAILED DISCUSSION OF MEMS
SWITCHES

Actuation Mechanisms: The actuation forces required for the mechanical movement can be obtained using electrostatic, magneto-static, piezoelectric or thermal designs. To date, only electrostatic-type switches have been demonstrated at 0.1-100GHz with high reliability at low RF powers for metal contact and medium power levels for capacitive contacts (100 4A1.4 Million to 50 Billion cycles depending on the manufacturer) and wafer-scale manufacturing techniques. Other switches which have demonstrated excellent performance are the Microlab Latching switch (up to 100 Million cycles) using magnetic actuation, and the thermal switches developed independently by Cronos Microsystems and the Univ. of California, Davis. It
is hard to test thermal switches for long cycle times


Switching Time: Electrostatic switches can be made small and with a very fast switching time (2-30 μs) while thermal/magnetic actuation requires around 100-2, 000 μs of switching time. An excellent metal-contact switch developed by LETI using thermal actuation but with an electrostatic hold, thereby requiring very little switching energy and virtually zero hold-down power. However, its switching time is still relatively slow (300 μs). The LETI switch has been tested to more than 100 million cycles. Contact Type: There are two different contacts in RF MEMS switches, a capacitive contact and a metalto- metal (or DC) contact. The capacitive contact is characterized by the capacitance ratio between the up-state (open circuit) and down-state (short-circuit) positions, and this is typically 80-160 depending on the design. The down-state capacitance is typically 2-3 pF, and is suitable for 8-100GHz applications. In general, it is hard to obtain a large down-state capacitance using nitride or oxide layers, and this limits the low-frequency operation of the device. On the other hand, DC-contact switches with small up-state capacitances (open circuit) can operate from 0.01 to 40GHz, and in some cases, to 60GHz (for example, the Rockwell Scientific switch has an up-state capacitance of only 1.75 fF and an isolation of 23 dB at 60GHz). In the down-state position (short-circuit), the DC-contact switch becomes a series resistor with a resistance of 0.5-2 Ω, depending on the contact metalused. Circuit and Substrate Configurations: As is the case with all two-terminal devices, the switches can be placed in series or in shunt across a transmission line. Typically, capacitive switches have been used in a shunt configuration, while DC-contact switches are placed in series. The reason is that it is easier to get a good isolation with a limited impedance ratio (such as the capacitive switch) in a shunt-circuit than in a series circuit. Also, MEMS switches are compatible with both microstrip and CPW lines on glass, silicon and GaAs substrates, and have been used in these configurations all the way to 100GHz. For low loss applications at microwave frequencies, it is important to use high-resistivity substrates.
CIRCUITS WITH RF MEMS SWITCHES
The near-ideal electrical response of RF MEMS witches (both metal-contact and capacitive) have allowed many designers to build state-of-the-art switching circuits from 0.1GHz all the way to 120GHz. In the past 4 years, these applications concentrated on the replacement of GaAs phase shifters which are commonly used in phased arrays by the thousands of units. A comparison between 3-bit GaAs phase shifters and MEMS phase shifters is shown in Table I and it is seen that MEMS switches provide an immense performance benefit especially at Ka-Band to W-band applications.




Fig. 4 presents a 4-bit miniature RF MEMS phase shifter developed jointly by the Univ. of Michigan and Rockwell Scientific. It is based on the Rockwell metalcontact switch and on CLC delay lines for miniaturization. The phase shifter results in an average loss of 1.4dB at 10GHz, a ±3◦ phase error, and is matched to −13 dB at the input and output ports from 6-16GHz. This phase shifter represents the smaller design using RF MEMS to-date, and with excellent response. Fig. 5 presents an 885-986MHz 5-pole tunable
filter using switched MEMS capacitors developed
by Raytheon Systems Co. In this case, capacitive switches are used to switch fixed-value metalinsulator- metal capacitors in the transmission line. The filter employs 18 switches and is a very complicated circuit with variable resonators and impedance inverters. Its measured response is nearly ideal, with excellent frequency tuning capabilities, very high linearity (in terms of measured IIP3) and a loss of 5- 6 dB due to the finite Q of the planar inductors used (Q = 30 at 0.9GHz). Fig. 6 presents a W-band 3-bit phase shifter developed at the Univ. of Michigan using MEMS capacitive switches [3]. This is the highest frequency MEMS phase shifter to-date and results in an average loss of 2.7-2.9 dB at 77-94GHz with an associated phase error of ±3◦. The results are about 8 dB better than GaAs designs.

Other circuits, which are not shown due to space constraints, are very wideband SP4T switches, highisolation series/shunt switches covering 0.1-50GHz, double-pole double-throw transfer switches, and a whole range of phase shifters from 8GHz to 120GHz. Also, tunable filters covering 200MHz to 23GHz have been developed by various groups. In general, RF MEMS circuits outperform GaAs FET and PIN diode circuits by a large margin at all frequencies of interest



the RF and microwave communities. Most of the
circuits developed in the world can be found in [1].
THE FUTURE
It is now clear that we understand RF MEMS switches well, both from the mechanical and electrical/ electromagnetic point of view. We can design complicated circuits using MEMS switches or varactors, and we can accurately predict their performance all the way to 120 GHz. They are still not accepted in the commercial and defense arena due to their need of a hermetic package, and their reliability under medium to high-power conditions. There is currently an intense effort to solve these problems, and the author
believes that RF MEMS switches and varactors will play an essential role in future high-value commercial and defense systems.
Jorge L Polentino U
19769972
CRF

the True Time Delay Phased Antenna Array

Beam Steering

The fundamental principles underlying the concept of electronic beam steering are derived from electromagnetic radiation theory employing constructive and destructive interference. These principles can be stated as follows: The electromagnetic energy received at a point in space from two or more closely spaced radiating elements is a maximum when the energy from each radiating element arrives at the point in phase. Controlling the phase through the many segments of the antenna system allows the beam to be rapidly directed in different directions. Figure 1 shows a 4-element linear array of antennas with constant phase difference between neighboring phase shifters, where θ0 is the scan angle, w1 to w4 are amplitude weights, and d is the spacing between adjacent antenna elements.

The angle θ0 of the beam with respect to the antenna axis is determined by the operating wavelength of the microwave signal, the spacing between the antenna elements that is usually half of a wavelength, and the phase shift between the signals in the individual elements. It is given by

(1)
where l is wavelength, d is the inter-element array spacing, nd is the location of the particular radiating element n being investigated, and f is the phase shift located at the nth element. It is seen that changing the frequency results in a change in steering angle q0, if the phase shift f is fixed. Therefore, the beam squinting arises from this distortion.


Instead, the group time delay t only depends on beam position angle and array length, but not frequency, which is shown in Eq. (2).







Beam shaping



Although steering capability is the most common function, a phased antenna array can also provide beam-shaping capability by appropriate arrangement of the feed signals. A radar system with variable beamwidth can produce a wide beam for the acquisition of targets and a narrow beam for subsequent high-precision tracking. Additionally, a broadcast satellite antenna with variable beamwidth can achieve efficient coverage of irregularly shaped geographical service areas based on the environment or traffic conditions, which is an important feature for communication systems.






The major objectives of beam shaping are to minimize pattern ripples, to reduce sidelobe levels, change null positions, or control output power levels, etc. To vary the antenna beamwidth, there are many methods which may be employed. In this research work, phase-only adjustment is used to vary the beamwidth. Adding a quadratic phase error on the array aperture moves the array’s phase center and changes its focusing distance (see Fig. 2). However, for a given size of an array aperture, there is a limit to the movement of the phase center. Beyond a certain value of phase taper across the array aperture, beam bifurcation on the plane of the linear array occurs. In addition, the sidelobe level of an array with equal power feeding increases rapidly as the value of phase taper increases. In order to increase the useful range and control the sidelobe level, a Taylor N-bar amplitude taper is introduced across the array aperture in this design.




















True Time Delay Technologies




The bandwidth of a phased antenna array is affected by many factors, including change of element input impedance with frequency, change of array spacing in wavelengths that may allow grating lobes, change in element beamwidth, etc. When an array is scanned with fixed values of phase shift, provided by phase shifters, there is also a bandwidth limitation as the position of the main beam will change with frequency, which is called beam squint. In Eq.1, It is obvious that changing the frequency results in a change of the scan angle for fixed element spacing. In contrast, when the array is scanned with true time delay, the beam position is independent of frequency to first order.Three representative true time delay (TTD) technologies are being investigated in our group. They are:


· Piezoelectric-Bender Controlled Delay Line;

· RF-MEMS Extended Tuning Range Varactor Delay Line;

· Liquid Crystal Phase Shifter



Jorge L Polentino U

19769972

CRF

viernes, 28 de mayo de 2010

Broadband Microstrip Patch Antenna

With the ever-increasing need for mobile communication and the emergence of many systems, it isimportant to design broadband antennas to cover a wide frequency range. The design of an efficientwide band small size antenna, for recent wireless applications, is a major challenge. Microstrip patchantennas have found extensive application in wireless communication system owing to theiradvantages such as low-profile, conformability, low-cost fabrication and ease of integration with feednetworks(He et al., 2008). However, conventional microstrip patch antenna suffers from very narrowBroadband Microstrip Patch Antenna 175bandwidth, typically about 5% bandwidth with respect to the center frequency. This poses a designchallenge for the microstrip antenna designer to meet the broadband techniques (Lau et al., 2006,Zhang & Wang, 2006).There are numerous and well-known methods to increase the bandwidth of antennas, includingincrease of the substrate thickness, the use of a low dielectric substrate, the use of various impedancematching and feeding techniques, the use of multiple resonators, and the use of slot antenna geometry(Pozer, 1995,Wi et al., 2002, Wi et al., 2006, Matin et al., 2007). However, the bandwidth and the sizeof an antenna are generally mutually conflicting properties, that is, improvement of one of thecharacteristics normally results in degradation of the other.Recently, several techniques have been proposed to enhance the bandwidth. A novel singlelayer wide-band rectangular patch antenna with achievable impedance bandwidth of greater than 20%has been demonstrated (Yang et al. 2001). Utilizing the shorting pins or shorting walls on the unequalarms of a U-shaped patch, U-slot patch, or L-probe feed patch antennas, wideband and dual-bandimpedance bandwidth have been achieved with electrically small size in (Guo et al., 2002, Chair et al.,2005). Other techniques involves employing multilayer structures with parasitic patches of variousgeometries such as E, V and H shapes, which excites multiple resonant modes. However, theseantennas are generally fabricated on thicker substrates (Bao & Ammann, 2007).In this paper, a novel slotted shape patch is investigated for enhancing the impedancebandwidth on a thin substrate (about 0.01 0λ ). The design employs contemporary techniques namely,the L-probe feeding, inverted patch, and slotted patch techniques to meet the design requirement. Inthis paper, the design and simulations results of the novel wideband microstrip patch antenna, isdescribed.Antenna Design and StructureIt is known that increasing the thickness of the patch antenna will increase the impedance bandwidth.However, the thicker the substrate of the antenna, the longer the coaxial probe will be used and, thus,more probe inductance will be introduced (Yang et al. 2001), which limits the impedance bandwidth.Consequently, a patch antenna design that can counteract or reduce the probe inductance will enlargethe impedance bandwidth.Figure 1 depicts the geometry of the proposed patch antenna. The inverted rectangular patch,with width W and length L is supported by a low dielectric superstrate with dielectric permittivity ε1and thickness h1. An air-filled substrate with dielectric permittivity εo and thickness ho is sandwichedbetween the superstrate and a ground plane.The proposed patch integrates both the E- and H-shaped patch on the same radiating element.For the E-shaped, the slots are embedded in parallel on the radiating edge of the patch symmetricallywith respect to the centerline (x-axis) of the patch and for the H-shaped the slots are embedded in serialon the non-radiating edge of the patch. The E- and H-shaped are shown in Figure1(a), where, l and ware the length and width of the slots. The patch is fed by an L-shaped probe with height, hP andhorizontal length, lP along the centerline (x-axis) at a distance fP from the edge of the patch as shown inFigure 1(b). Table 1 shows the optimized design parameters obtained for the proposed patch antenna.A Rogers RT 5880 Duroid™ dielectric substrate with dielectric permittivity, ε1 of 2.2 and thickness, h1of 0.01 0 λ has been used in this research. The thickness of the air-filled substrate, ho is 16.5 mm. AnAluminum plate with dimensions of 1.378 0 λ ×1.23 0 λ and thickness of 1 mm is used as the groundplane. The proposed antenna is designed to operate in the 1.88 GHz to 2.22 GHz region. The use of Lprobefeeding technique with a thick air-filled substrate provides the bandwidth enhancement, whilethe application of superstrate with inverted radiating patch offers a gain enhancement, and the use ofparallel and series slots reduce the size of the patch. The use of superstrate on the other hand wouldalso provide the necessary protections for the patch from the environmental effects. By incorporatingextra slots in radiating edges, the gain and cross polarization has been improved. In addition, the176 Mohammad Tariqul Islam, Mohammed Nazmus Shakib, Norbahiah Misran and Tiang Sew Sunantenna is relatively compact in comparison with the slotted antenna described in (Tariqul Islam et al.,2007). These techniques offer easy patch fabrications, especially for array structures







Simulation Setup

The resonant properties of the proposed antenna have been predicted and optimized using a frequencydomain three- dimensional full wave electromagnetic field solver (Ansoft HFSS). The design flowdiagram is shown in Figure 2, starting with a baseline design of the inverted rectangular patch with anBroadband Microstrip Patch Antenna air-filled dielectric, the baseline parameters (L, W, h1, and ho) are determined at centre frequency, fo anddielectric permittivity ε1. The H- and double E-shaped are then introduced on the patch with the initialvalues slots parameters to reduce the patch size and crosspolarization level. Next, the L-probe isintroduced to feed the patch and its parameters are adjusted to achieve the broadband requirement.





Results and Discussion
shows the simulated result of the return loss of the proposed antenna. The two closely excitedresonant frequencies at 1.93 GHz and at 2.18 GHz as shown in the figure gives the measure of thewideband characteristic of the patch antenna. The simulated impedance bandwidth (VSWR≤2) of21.79% from 1.84 GHz to 2.29 GHz is achieved at 10 dB return loss.Mohammad Tariqul Islam, Mohammed Nazmus Shakib, Norbahiah Misran and Tiang Sew Sun

The simulated radiation patterns at the second resonant frequency in the xz-plane and yz-planeare plotted in Figure 4. For the sake of brevity, only calculated radiation pattern for second resonancefrequency is given in this paper. As shown in Figure 4, the designed antenna displays good broadsideradiation patterns in the xz-plane and yz-plane at upper resonant frequency. It can be seen that 3-dBbeamwidth of 65º and 50º for xz-plane and yz-plane respectively at 2.18 GHz. The crosspolarizationpattern is lower than about -35dB in xz-plane. The L-probe inverted proposed patch antenna exhibitsbetter crosspolarization than the design reported in (Tariqul Islam et al., 2007). Notable, the radiationcharacteristics of the proposed antenna are better to those of the conventional patch antenna. Theradiation patterns at other bands, which are similar to those at 1.93 GHz, are not presented here indetail.




The simulated gain of the L-probe fed inverted proposed patch antenna at various frequencies isshown in Figure 5. As shown in the figure, the maximum achievable gain is 9.5 dBi at the frequency ofBroadband Microstrip Patch Antenna 1792.08GHz. The design has better gain variation of 0.9dBi compared to conventional rectangular patchdue to the embedded slots.

Conclusion
A novel technique for enhancing bandwidth of microstrip patch antenna is successfully designed in thisresearch. Simulation results of a wideband microstrip patch antenna covering 1.885 to 2.200 GHzfrequency have been presented. Techniques for microstrip broadbanding, size reduction, andcrosspolarization reduction are applied with significant improvement in the design by employingproposed slotted patch shaped design, inverted patch, and L-probe feeding.The proposed microstrip patch antenna achieves a fractional bandwidth of 21.79% (1.84 to 2.29GHz) at 10 dB return loss. The maximum achievable gain of the antenna is 9.5 dBi with gain variationof 0.9dB. The proposed patch has a compact dimension of 0.544 0λ × 0.275 0λ .The wideband characteristic of the antenna is achieved by using the L-shaped probe feedingtechniques, the use of series slots (H-shaped) and use of another pair of parallel slots (E-shaped) lead tothe patch size reduction. Better radiation performance is achieved by embedding parallel slots onto thepatch (E-shaped) while the use of inverted patch improves the gain of the antenna. The compositeeffect of integrating these techniques offer a low profile, broadband, high gain, and compact antennaelement suitable for array applications.
Jorge L. Polentino U.
19769972
CRF

jueves, 27 de mayo de 2010

Antena Dipolo de media onda

Una forma fundamental de antena es un único hilo cuya longitud sea aproximadamente igual a un medio de la longitud de onda de transmisión. Es la unidad a partir de la que se construyen muchas de las formas más complejas de antenas y se conoce como la antena dipolo.
La longitud de media onda en el espacio libre es
Longitud (pies)= 492/ f(MHz) (Ec. 1)
Longitud (metros)= 150/ f(MHz )
La longitud real de una antena de media onda resonante no será exactamente igual a media longitud de onda en el espacio, sino que depende del tamaño del conductor respecto a la longitud de onda. La relacion se muestra en la figura 3, en la que K es un factor que debe multiplicarse por media longitud de onda en el espacio libre para obtener la longitud de la antena resonante. En las antenas de hilo sujetas mediante aisladores en los extremos se produce un efecto adicional que acorta la longitud debido a la capacidad añadida al sistema por los aisladores (efecto de puntas). La siguiente fórmula es lo bastante precisa para antenas de hilo para frecuencias hasta 30 MHz:
Longitud de la antena de media onda (pies) =
=492 x 0,95/ f(MHz)= 468/ f(MHz) (Ec. 2)
Longitud de la antena de media onda (metros)
=142,5/f(MHz)
Ejemplo: Una antena de media onda para 7150 kHz (7,15 MHz) tiene 468/7,15 = 64,45 pies o 65 pies 5 pulgadas (19,93 m).
Por encima de 30 MHz deben utilizarse las siguientes fórmulas, especialmente para antenas construidas con barra o tubo. K se toma de la figura 3.
Longitud de la antena de media onda (pies) =
492 x K /f(MHz) (Ec. 3)
Longitud (pulgadas)=5904 x K/ f(MHz) (Ec. 4)
L(metros) =150 x K/f(MHz)
L(centímetros) = 15000 x K /f(MHz)
Ejemplo: Determinar la longitud de una antena de media onda en 28,7 MHz, si la antena está construida con tubo de media pulgada de diámetro, En 28,7 MHz, media longitud de onda en el espacio libre son:
492/28,7 = 17,14 pies
utilizando la ecuación 1. La relación de media longitud de onda al diámetro del conductor (cambiando la longitud de onda a pulgadas) es:
(17,14 x12)/ 0,5 = 411
De la figura 3, K= 0,97 para esta relacion, La longitud de la antena a partir de la ecuación 3 es:
492 x 0,97/28,7 = 16,63 pies
o 16 pies 7-1/2 pulgadas. La solución se obtiene directamente en pulgadas sustituyendo en la ecuación 4:
5904 x 0,97/28,7 = 199,5 pulgadas
La longitud de una antena de media longitud de onda también está afectada por la proximidad de los extremos del dipolo a objetos conductores o semiconductores cercanos. En la práctica, a menudo es necesario hacer un ligero "recorte" experimental del hilo después de cortar la antena a la longitud calculada, alargando o acortando en pequeños incrementos para obtener una ROE baja. Esto puede hacerse aplicando potencia de RF a través de un medidor de ROE y observando la lectura de potencia reflejada. Cuando se obtiene la ROE más baja para el segmento de la banda de radioaficionado deseada, la antena es resonante a esa frecuencia. El valor de la ROE indica la calidad de la adaptación entre la antena y la línea de alimentación. Con las impedancias de línea de alimentación de 50 a 75 ohms, la ROE en resonancia debería estar entre 1: 1 y 1,7:1 para antenas a una altura media y despejada respecto a objetos conductores próximos. Si la ROE más baja que pueda obtenerse es demasiado alta para utilizarla con equipos de estado sólido, puede utilizarse un Transmatch o red de acoplamiento de entrada a la línea.
Características de radiación
El clásico diagrama de radiación de una antena dipolo es más intenso perpendicular al hilo. Si el dipolo está a una altura de media longitud de onda o más y no está degradado por objetos conductores próximos puede suponerse que el diagrama de radiación tiene forma de ocho. Esta suposición también se basa en una alimentación simétrica.
En la práctica, una línea de alimentación coaxial puede distorsionar ligeramente el diagrama, como se muestra en la figura 4.

El mínimo de radiación horizontal se produce hacia los extremos del dipolo. Esto se aplica a una antena de media onda que esté paralela al suelo. Sin embargo, si el dipolo está vertical se producirá una radiación uniforme en todas direcciones con una forma de rosquilla si se viera desde la parte superior de la antena. Muchos principiantes suponen que la antena dipolo presentará una gráfica así a cualquier altura sobre el suelo. De hecho, a medida que la antena se acerca al suelo, el diagrama de radiación se deteriora hasta que la antena es, en su mayor parte, un radiador omnidireccional de ondas de angulo alto. Muchos han intentado utilizar cualquier altura conveniente, como 20 o 30 pies sobre el suelo, para un dipolo de 80 metros, sólo para comprobar que el sistema es efectivo en todas direcciones para distancias relativamente cortas (500 y 1000 millas en buenas condiciones). De todo esto puede deducirse que la altura sobre el suelo es importante por multitud de razones. La figura 5 ilustra claramente las ventajas que se obtienen con la altura de la antena.









El ángulo de radiación de la figura 5A es de 300, mientras que a una altura de una longitud de onda (fig. 5B) el lóbulo se desdobla y proporciona un buen ángulo para comunicaciones DX de 15 grados. La directividad de la antena a las dos alturas se muestra en la figura 5C. La línea continua muestra el diagrama de azimut para un ángulo de radiación de 30 grados de un dipolo a 1/2 onda de altura y corresponde a la gráfica de la figura 5A. La línea de puntos muestra el diagrama para un ángulo de radiación de 15 grados y 1 onda de altura, correspondiente a la gráfica de la figura 5B. La figura 5C ilustra que hay una importante radiación en los extremos de un dipolo horizontal bajol incluso en el ángulo de elevación más favorable.


Para la altura de ½ onda (línea continua), la radiación por los extremos sólo es 7,6 db más baja que en la dirección perpendicular. Los lóbulos de ángulo alto del diagrama de la figura 5B (50 grados) son útiles para comunicaciones por salto corto y, en la práctica, se comparan favorablemente con el lóbulo mostrado en la figura 5A. A alturas apreciablemente menores de 1/2 longitud de onda, el lóbulo se hace más alto. Finalmente, los dos lóbulos convergen para formar una "bola de radiación" única con ángulos muy altos (deficiente para comunicaciones de larga distancia).





Métodos de alimentación
La mayoría de antenas dipolo de hilo de aficionado (media longitud de onda) tienen una impedancia en el punto de alimentación entre 50 y 75 ohms, dependiendo de la instalación. Por tanto, el cable coaxial normalizado es adecuado para la mayoría de instalaciones. Los cables de los tipos más pequeños (RG-58/U y RG-59/U) son satisfactorios con niveles de potencia hasta algunos cientos de vatios si la ROE del sistema es baja. Para las estaciones de alta potencia deben emplearse los cables muy gruesos (RG-8/U y RG-11/U). Estos cables pueden conectarse al centro de la antena, como se muestra en la figura 6.












Se utiliza un bloque aislante de plástico como refuerzo central para el cable y los hilos del dipolo. La malla del coaxial se conecta a uno de los hilos y el conductor central se suelda a la otra rama. El extremo abierto del cable debe protegerse contra el polvo y la polución para evitar la degradación de la línea de transmisión.





Puede obtenerse alimentación simétrica colocando un transformador balun de 1:1 en el punto de alimentación del dipolo. Si no se utiliza, es improbable que se aprecie el ligero desplazamiento del diagrama de radiación resultante de la alimentación asimétrica. Los efectos de una alimentación asimétrica son más significativos en antenas directivas en VHF y superiores. Cuanto más estrecho sea el diagrama de radiación de la directiva, más molesta será la situación. La impedancia característica de una antena dipolo puede aumentarse utilizando un dipolo de dos hilos o dipolo plegado, del tipo mostrado en la figura 7.










Esta antena ofrece una buena adaptación para la línea plana de TV de 300 ohms.



Alternativamente, podrían utilizarse dos hilos para formar un equivalente de dipolo de la línea de TV. Si se hace esto será necesario colocar espaciadores aislantes cada pocos pies a lo largo de todo el dipolo para mantener separados uniformente los hilos del dipolo y para evitar el cortocircuito. La línea en escalera de hilos abiertos de TV es excelente para utilizarla en una antena dipolo plegado de 300 ohms, tanto para el radiador como para la línea de alimentación. Las pérdidas en el alimentador serán muy bajas con este tipo de construcción comparándolas con las del cable de hilos paralelos moldeado de TV. La antena dipolo puede utilizarse como un radiador toda banda empleando una línea de alimentación abierta sintonizada









En ese ejemplo el dipolo se corta a media longitud de onda para la banda de aficionado más baja que se desee. Funciona en sus armónicos cuando se utiliza en las otras bandas de aficionado seleccionadas. Una antena típica de este tipo podría utilizarse de 80 a 10 m. Esta forma de radiador es conocida por algunos aficionados como Zeppelin alimentado en el centro. En la figura 8B se muestra una versión con alimentación por el extremo (Zepp alimentado por punta).



Esta versión no es tan adecuada como la versión alimentada en el centro debido a que el sistema de alimentación no es simétrico. Esto puede producir radiación del alimentador y una distorsión del diagrama de radiación de la antena. Ambos tipos de antena Zepp precisan de una red de adaptación (Transmatch) en el extremo del transmisor de la línea para convertir la impedancia del alimentador a 50 ohms y para cambiar la condición equilibrada a una no equilibrada.


Aunque la línea de alimentación puede ser cualquiera desde 200 hasta 600 ohms




(no es crítico), las pérdidas serán insignificantes cuando se utilizan líneas abiertas.


Esto es cierto independientemente de las variaciones de la impedancia del punto de alimentación del dipolo de banda a banda. La impedancia del punto de alimentación será alta en los armónicos pares y baja en la frecuencia de funcionamiento más baja y en sus armónicos impares. Por ejemplo, si se corta el dipolo para 40 m, la impedancia del punto de alimentación será baja en 40 y 15 m, pero será alta en 20 y 10 m. Con una antena dipolo la línea de alimentación debe alejarse de la antena en ángulo recto hasta tanta distancia como sea posible. Esto ayudará a evitar el desequilibrio de corrientes en la línea producido por la captación de la RF del dipolo. Se recomienda un alejamiento en ángulo recto de 1/4 de longitud de onda o más. En algunas circunstancias puede ser necesario experimentar con la longitud del alimentador de línea abierta cuando se utiliza un Zepp toda banda. Esto se debe a que en ciertas frecuencias de funcionamiento la línea puede presentar una impedancia "dificil" al Transmatch, haciendo imposible obtener una condición de carga adecuada para el transmisor. Esto dependerá de la capacidad del Transmatch que se utilice.


Jorge L. Polentino U.
19769972
CRF